Layout technique for matched resistors on an integrated circuit substrate

ABSTRACT

Provided a method of reducing impedance variations in an electrical circuit structured and arranged for placement on an integrated circuit (IC) substrate. The method includes forming sets of parallel connected resistors, each set corresponding to one of the impedance devices on the IC. Each set also includes two or more parallel resistor paths, each resistor path including two or more cascaded resistors and has a total impedance value substantially equal to the predetermined impedance value of its corresponding impedance device. Finally, the method includes configuring the sets of parallel resistor paths to form an interdigital structure across the substrate when the electrical circuit is placed on the IC.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional Application No.60/350,035, filed Jan. 23, 2002, entitled “System and Method for aProgrammable Gain Amplifier,” which is incorporated by reference hereinin its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to the field of low-noiseamplifiers. In particular, the present invention relates to thedevelopment of low-noise programmable gain amplifiers (PGAs) suitablefor placement on integrated circuits (ICs) and for use in signalprocessing applications.

2. Related Art

PGAs are used in various analog signal processing applications where anelectrical signal of varying amplitude must be either amplified orattenuated before subsequent signal processing. Various gain and/orattenuation settings are required to accommodate the wide dynamic rangeneeded for the amplifier's input stages. Numerous conventionaltechniques exist for meeting these demands.

What are needed, however, are techniques for providing attenuation inclosed loop amplifiers without increasing their feedback factor. What isalso needed is an approach to ensure suitable start-up conditions andavoid latchup, particularly in complimentary metal oxide semiconductor(CMOS) PGAs. Finally, a technique is needed to eliminate mismatchedcharacteristics commonly found in passive elements across IC substratesdue to process gradients.

SUMMARY OF THE INVENTION

The present invention includes a method and system for reducingimpedance variations in an electrical circuit structured and arrangedfor placement on an integrated circuit (IC) substrate. The methodincludes forming sets of parallel connected resistors, each setcorresponding to one of the impedance devices on the IC. Each set alsoincludes two or more parallel resistor paths, each resistor pathincluding two or more cascaded resistors and has a total impedance valuesubstantially equal to the predetermined impedance value of itscorresponding impedance device. Finally, the method includes configuringthe sets of parallel resistor paths to form an interdigital structureacross the substrate when the electrical circuit is placed on the IC.

BRIEF DESCRIPTION OF THE FIGURES

The accompanying drawings, which are incorporated in and constitute partof the specification, illustrate embodiments of the invention and,together with the general description given above and detaileddescription of the embodiments given below, serve to explain theprinciples of the present invention.

FIG. 1 is an illustration of a conventional sub-PGA module having a highinput resistance;

FIG. 2 is an illustration of a standard PGA module with activeattenuation;

FIG. 3 is a block diagram illustration of an exemplary receive pathconstructed and arranged in accordance with the present invention;

FIG. 4 is a more detailed block diagram illustration of a first stagePGA shown in FIG. 3 constructed and arranged in accordance with thepresent invention;

FIG. 4A is a block diagram illustration used to depict the relationshipbetween impedance, gain, and feedback factor within the first stage PGAof FIG. 4.

FIG. 5 is a schematic diagram of a conventional common mode feedbackcircuit which can be used to insure proper start-up conditions in thePGA of FIG. 4;

FIG. 6 is a block diagram of the amplifier in the PGA of FIG. 4,including an exemplary start-up circuit constructed and arranged inaccordance with the present invention;

FIG. 7 is a flow chart of a method of using the start-up circuit of FIG.6;

FIG. 8A is an illustration of a first step of a resistor layoutconstructed and arranged in accordance with the present invention;

FIG. 8B is an illustration of a second step of the resistor layout ofFIG. 8A; and

FIG. 8C is an illustration of a third step of the resistor layouts ofFIGS. 8A and 8B.

DETAILED DESCRIPTION OF THE INVENTION

The following detailed description of the accompanying drawingsillustrates exemplary embodiments consistent with the present invention.Other inventions are possible, and modifications may be made to theembodiments within the spirit and scope of the invention. Therefore, thefollowing detailed description is not meant to limit the invention.Rather, the scope of the invention is defined by the appended claims.

It would be apparent to one of skill in the art that the presentinvention, as described below, may be implemented in many differentembodiments of hardware, software, firmware and/or the entitiesillustrated in the figures. Any actual software code with thespecialized control hardware to implement the present invention, is notlimiting of the present invention. Thus, the operation and behavior ofthe present invention will be described with the understanding thatmodifications and variations of the embodiments are possible, given thelevel of detail presented herein.

FIG. 1 is an illustration of a conventional pre-attenuator 100 followedby a sub-PGA 102 with high-input resistance. In this topology, thepre-attenuator 100 is a separate passive attenuator followed by thesub-PGA 102, which has a high input resistance. A sub-PGA is a circuitblock that can provide a variable gain, but does not necessarily provideattenuation. The sub-PGA 102 can consist of one or more circuit blocks.In particular, the sub-PGA 102 may contain a buffer followed by a moretraditional PGA module.

The key characteristic of the sub-PGA 102 is that it typically has highinput resistance. Because of the high input resistance of the sub-PGA102, the pre-attenuator 100 and sub-PGA 102 do not interact, thusreducing the possibility of a beneficial reduction in the sub-PGA 102'sfeedback factor. The feedback factor is a numerical index that isrelated to the ratio of the resistors in closed loop amplifiers thathave resistive feedback networks. The ratio of the resistors changes thefeedback factor which in-turn changes the amplifier's gain. Whenamplifier's are used to provide attenuation, the feedback factortypically increases, which consequently makes the amplifier'sperformance unstable. Therefore, the relationship between gain and theamplifier's feedback factor becomes problematic when the amplifier isused as an attenuator.

One alternative approach for providing attenuation in amplifiers is tohave a passive or an active programmable attenuator followed by aseparate, programmable gain amplifier. If the programmable attenuatorand the PGA are separate, however, a feedback factor reduction will bedifficult to achieve. Another alternative approach is to use theprogrammable gain amplifier as an active attenuator. However, using theprogrammable gain amplifier as an active attenuator creates twosignificant problems. First, the issue of the feedback factor becomes amore significant consideration, and the associated substrate arearequired to accommodate the amplifier and the attenuator becomeextremely large.

Next, a standard PGA circuit 200 with active attenuation is shown inFIG. 2. In the topology of the PGA circuit 200, there is a standardinverting-gain PGA 202 where attenuation is achieved in the feedbacknetwork by making an input resistance 204 larger than a feedbackresistance 206.

When using CMOS devices, there is a need to guarantee that the PGAcircuits such as the PGA circuit 200, start up and achieve desirableoperating points under a wide variety of start-up, bias, andenvironmental conditions. In 2-stage differential amplifiers that aretypically associated with CMOS circuits, the common-mode (CM) signalloop of the first stage amplifier has net positive feedback. Therefore,it is possible that this positive feedback may cause the amplifier toreside in an unwanted latch-up state, especially if the amplifier'spositive CM feedback characteristics are able to overpower the negativeCM feedback characteristics of the second stage amplifier, also known asthe CM feedback amplifier (CMFBA).

Conventional approaches for eliminating the start-up problem includemaking the CMFBA large enough so that the primary amp CM loop (positivefeedback) can never overpower the CMFBA loop (negative feedback). Thisapproach might be considered a high-power, high-noise solution.

Another conventional approach for resolving the start-up problemincludes providing a high impedance value pull-up resistor to forcevoltage on a common source (CM-src) node. For an amplifier configured inthis manner, a high-valued resistor connected from a supply voltagesource V_(DD) to the CM-src node can pull up the CM-src node in order toforce proper start-up. This approach, however, requires extra power andadds noise to the associated system.

Another challenge to the development of PGAs suitable for placement onICs is that the ICs often require passive elements, such as resistors,that are well-matched to one another. Silicon processing, for example,is an imperfect process that frequently results in process gradients,where the characteristics of a particular device will vary roughlylinearly across a certain dimension of the IC's substrate. This lineargradient can cause severe mismatch between devices, such as resistors,that need to be well-matched to one another.

In order to achieve a sufficient gain range and gain resolution (i.e.,gain step size), complex resistive feedback networks are typically usedwithin the amplification modules. This comes at the cost of increasedassociated die sizes. Additionally, since closed-loop gain and feedbackfactor are inversely related, in order to achieve a large gain range, itoften requires the feedback factor to span a wide range. Thiscomplicates the design of the amplifier, which must be guaranteed to bestable and functional over the wide feedback factor span. All theseproblems must be solved while keeping the closed-loop performance of thePGA sufficiently linear.

One aspect of the present invention merges a passive pre-attenuatornetwork with the feedback network used within the amplifier. PGAfunctionality is achieved by using a closed-loop amplifier with aswitchable resistor network in the feedback loop to provide passiveattenuation. This accomplishes three goals. Since the pre-attenuator canbe controlled separately from the rest of the feedback network, itallows for greater controllability of the net PGA gain. Therefore, thetotal complexity of the feedback network can be reduced.

Next, the nature of the pre-attenuator of the instant invention is suchthat when it's used to reduce the overall PGA gain, the overall feedbackfactor is reduced as well. By contrast, if a conventional feedbacknetwork is used to reduce the overall PGA gain, the overall feedbackfactor increases. Therefore, using the present invention and withcareful partitioning of the overall gain between the pre-attenuator andthe remainder of the feedback network, one can achieve a wide gain rangewith a much smaller variation in the feedback factor. This aspect easesthe circuit design of the amplifier.

Finally, the first stage feedback network is configured to have CMOSswitches placed on a virtual ground. Therefore, no signal current flowsthrough these switches when CMOS technology is used. Also, the switchesin the attenautor network have a symmetric differential drive applied tothem. Therefore, the linearity of the network will be sufficiently high.

FIG. 3 provides an illustration of an exemplary PGA 300, its analogfront end, and associated digital circuitry. The general purpose of thePGA 300 is to provide gain (amplify) an input signal having a smallamplitude so that downstream analog-to-digital converters (ADCs) canreceive it and sample a signal having a sufficiently large amplitude.Therefore, the PGA is the first block positioned along a receive path302 of the PGA 300. Although any number of PGA stages can beaccommodated, for purposes of illustration, the PGA 300 includes threeamplification stages 308, 310, and 312.

The PGA 300 of FIG. 3 can also include an input buffer 314 for driving aswitched capacitor circuit (not shown) inside an ADC 315. Next, a lowpass filter 316 is included for removing unwanted energy at out-of-bandfrequencies. The input buffer 314 is a fourth amplifier separate andapart from the three amplification stages 308, 310, and 312 of the PGA300. A first aspect of the PGA 300 is associated with switching insideof the PGA, pre-attenuation characteristics, and providing a lowerfeedback factor.

FIG. 4 is a more detailed illustration of the first amplification stage308 of the PGA 300 shown in FIG. 3. Particularly, FIG. 4 shows aresistive feedback network 400 including network segments 401-403(discussed in greater detail below) connected with an amplifier 404. Inthe exemplary embodiment of FIG. 4, the amplifier 404 is a differentialamplifier including differential input and output ports. The networksegments 401 and 402 are structurally identical. Since the segments 401and 402 are attached to respective symmetrical differential portions ofthe amplifier 404, they are also functionally identical.

A dotted line 405 indicates an axis of symmetry regarding the layout ofthe amplifier 404. Therefore, comments directed to the network segment401 will also apply to the network segment 402. Additionally, thenetwork segment 403 is functionally structured along the amplifier'ssymmetrical axis 405. The voltage applied to the network segments401-403 is also symmetric and differential. This symmetric differentialswing prevents any non-linearities that might be associated with theCMOS process from being excited. Therefore, the linearity of theamplifier 404 and the resistive network 400 will be sufficiently high.

Another important function of the PGA 300 is to provide an amplifierhaving a settable or programmable gain. Progammability can be achievedby altering the input and output impedances of the feedback network 400.Gain, for example, is a function of the ratio of resistor impedanceswithin the network segments 401-403. The network segments 401-403 shownin FIG. 4 are connected between non-inverting amplifier input port Vg+and inverting output port Vo− and between inverting input port Vg− andnon-inverting output port Vo+ of the amplifier 404. In the process ofchanging the impedance of the resistors within the network segments401-403, the feedback factor and the gain of the amplifier 404,correspondingly change.

Changing the feedback factor has a number of effects concerning issuessuch as the linearity of an amplifier, the noise of the amplifier, andthe amplifier's stability. The higher the feedback factor, the betterthe performance of the amplifier in terms of noise and linearity.However, the amplifier may lose a measure of stability if the feedbackfactor is permitted to get too high. In other words, it's more difficultto maintain the stability of the amplifier if its feedback factor is toohigh.

As noted above, the feedback factor is a numerical index derived basedupon the topology of the amplifier and the feedback network. It isrelated to the ratio of the resistor values within the amplifier'sassociated resistive feedback network. For example, given an amplifierwith a resistive feedback network, a user can directly calculate thefeedback factor, which will typically be a number between zero and one.The feedback factor is also known as the beta-factor of a closed loopamplifier and is more of a quantitative, as opposed to a qualitative,measure of the amplifier's performance. Therefore, changing the ratio ofthe resistor impedance values changes the feedback factor, which in-turnchanges the gain of the amplifier.

In traditional approaches, there is a one-to-one correspondence betweenthe desired gain and the feedback factor present in the amplifier. Theyare essentially inversely related. The terms are not directly inverselyrelated because the associated mathematical expressions depict a morecomplicated relationship.

If one desires a low gain or equivalently, desires attenuation, thefeedback factor of conventional amplifiers increases and asymptoticallyapproaches the value one, making it harder for the amplifier to becomestable.

On the other hand, if one desires a higher gain, the feedback factorreduces and asymptotically approaches the value of zero, and it becomeseasier to make the amplifier stable. Conventional first stage amplifiershave a wide spread in terms of gain. For example, ranges of about −12 dBto +12 dB are not uncommon and are representative of voltage gains ofabout 0.25 to 4. Thus, the feedback factor can vary by a substantialamount in these conventional amplifiers.

In the present invention, however, the resistive network segment 403 isconfigured to cooperate with the resistive network segments 401 and 402to facilitate small gains in the amplifier 404 without increasing itsfeedback factor. More specifically, the network segments 401 and 402facilitate traditional gain increases within the amplifier 404. Forexample, resistors 406 a 0, 406 a 1-406 an, and switches 406 b 1-406 bnof the network segment 401, cooperatively function to provide theamplifier 404 with gain values greater than or equal to one. Theresistor 406 a 0 is a first portion of the network segment 401, whilethe resistors 406 a 1-406 an and switches 406 b 1-406 bn collectivelyrepresent a second portion of the network segment 401.

The exemplary network segment 403, on the other hand, facilitates gainvalues of less than one. In particular, the network segment 403 enablesthe amplifier 404 to attenuate an input signal received at an input portVin+ of the amplification stage 308 without increasing the amplifier'sfeedback factor. When attenuation is desired, switches 408 b 1-408 bmcan be closed. The switches 408 b 1-408 bm provide attenuation for theamplifier 404 and simultaneously lower the feedback factor, consequentlyimproving the amplifier's stability. In other words, the switches 408 b1-408 bm form a pre-attenuator that enable the amplifier 404 to producegain values of less than one while also lowering its feedback factor.

Conversely, the: approach of the present exemplary embodiment of FIG. 4makes it easier to design amplifiers without concern for stabilityrelated issues. The presence of resistors 408 a 1-408 am both attenuatesthe input signal and lowers the amplifier's feedback factor. As notedabove, conventional closed loop amplifiers are configured such that theamplifier's attenuation ability and its feedback factor typicallyoperate in opposite directions. In these conventional amplifiers, forexample, when an input signal is attenuated, the amplifier's feedbackfactor typically increases.

In the present invention, however, the presence of the switches 408 b1-408 bm and the corresponding resistors 408 a 1-408 am allow for gainsetting whereby these resistors both attenuate the signal andsimultaneously lower the feedback factor. This effect is due to themathematical relationships between the PGA gain and the resistor valuesin the feedback network. The gain and the feedback factor can bedirectly calculated based upon equations (1) and (2), assuming that thenet resistances of each branch of the feedback network are asillustrated in FIG. 4A. $\begin{matrix}{{Gain} = \frac{R4}{\left\lbrack {\left( {{R1}*{R2}} \right)/{R3}} \right\rbrack + {R1} + {R2}}} & (1) \\{{{Feedback}\quad {Factor}} = \frac{{R2} + \left\lbrack {\left( {{R1}*{R3}} \right)/\left( {{R1} + {R3}} \right)} \right\rbrack}{{R4} + {R2} + \left\lbrack {\left( {{R1}*{R3}} \right)/\left( {{R1} + {R3}} \right)} \right\rbrack}} & (2)\end{matrix}$

In FIG. 4A, it is apparent that the values of R1, R2, and R4 arecontrolled based upon which switch of the set 406 b 1-406 bn is closed.Also, it is apparent that the value of R3 is controlled by whichswitch(es) of the set 408 b 1-408 bm is/are closed. In general, as moreof the switches of the set 408 b 1-408 bm are closed, the value of theR3 in FIG. 4A will go down. Examining the equations (1) and (2), one candirectly calculate that reducing the value of R3 simultaneously lowersthe gain and the feedback factor.

Therefore, it can equivalently be stated that: by closing one or more ofthe set 408 b 1-408 bm, both the gain and the feedback factor arereduced.

The switches 406 b 1-406 bn permit the amplifier 404 to achieve a widevariety of gain settings. At any given time, exactly one switch of theset 406 b 1-406 bn will be closed. All other switches in this set willbe opened. By closing a different switch of this set 406 b 1-406 bn, theoverall gain of the PGA 308 will be impacted. In general, the closer theswitch is to the input nodes (Vin+ and Vin−), the higher the resultantgain of the PGA will be when that switch is closed. For example, ifswitch 406 b 1 were to be closed, that would result in a higher PGA gainthat if switch 406 b 4 were to be closed.

Similarly, selectively closing and opening the switches 408 b 1-408 bmfacilitates the achievement of gain values of less than 1 while alsolowering the feedback factor. The presence of the resistors 408 a 1-408am facilitates both attenuation of the input signal and lowering of thefeedback factor. The resistors 408 a 1-408 am on the opposite sides ofthe respective switches 408 b 1-408 bm are mirror images of each other.

Also, in the present invention, passive attenuation characteristics areinherently part of the structure of the amplifier 404 and the resistivenetwork segments 401-403 that form the amplifier's feedback network 400.Thus, the amplifier 404 and the network segments 401-403 are completelyintegrated in terms of their structure and function. That is, theimpedances associated, for example, with the switches 408 b 1-408 bm areelectrically coupled to a feedback path 409 of the amplifier 404. Theresult of this coupling is that an attenuation matrix formed within thefeedback network 400 cannot be analyzed separately from the gain aspectof the amplifier 404. More precisely, the network segment 403 is builtinto the structure of the amplifier 404 and is inherently part of thefeedback network 400.

FIG. 5 is an illustration of a conventional common mode feedback circuit500 used to insure proper start-up conditions. As previously noted,particularly in CMOS circuits, there is a need to guarantee that the PGAcircuit starts up and achieves a desirable operating point under numberof environmental conditions. The conventional circuit 500 is configuredto accommodate most common-mode excursions and fluctuations found undernormal operating conditions. In essence, the circuit 500 essentiallyoperates as a common-mode feedback circuit correcting typical start-updeficiencies that might impact an amplifier's performance. These typicalstart up deficiencies, however, are not severe enough to render theamplifier inoperable.

In the circuit 500, a differential output signal provided at the outputterminals Vo− and Vo+ of the amplifier 404 in FIG. 4 is received at theinput terminals Vo− and Vo+ of the circuit 500. A resistor string 501,connecting the input terminals Vo− and Vo+, averages the two voltages asVcmout so that it becomes the common-mode output of the amplifier.Vcmout is then compared with an internally generated reference voltageVcmref within differential pair transistors 502 and 504. If Vcmout ishigher than Vcmref, then the differential transistor pair 502/504 istilted so that more current flows through transistor 502 than transistor504. This pulls a common-mode feedback voltage Vcmfbp at a common-modefeedback transistor 506, low.

Thus, if the common-mode output is too high, the circuit 500 pull thecommon-mode feedback voltage Vcmfbp low. When Vcmfbp is pulled low, thenVo− and Vo+ are also pulled low via a correcting signal provided atcommon-mode outputs 506 and 508, which is in-turn provided as an inputto the amplifier 404. The circuit 500 is therefore effective atcorrecting minor start-up deficiencies in the amplifier 404 such asfluctuations in the common-mode voltage Vcmfbp. The circuit 500,however, has a somewhat limited range and capability. That is, althoughits effective against, for example, minor common-mode fluctuations, itis not effective at eliminating more severe start-up deficiencies suchas common-mode latch-up, which can render the amplifier 404 completelyinoperable.

FIG. 6 provides a more detailed illustration of the amplifier 404 shownin FIG. 4, including an exemplary start-up circuit 600 configured toeliminate the more severe start-up deficiencies such as common-modelatch-up, which can render the amplifier-404 inoperable.

In the illustrations of FIGS. 4 and 6, the input CM, the output CM, andthe CM of the gate of the amplifier 404 are significant structuralfactors. That is, the input to the resistor network, the input to theamplifier, and the output to the amplifier are all dependent upon oneanother. There are no additional DC paths connected to the input of thePGA 308 (nodes Vin+ and Vin− of FIG. 4). Therefore, from the DCstandpoint, the input to the PGA 308 is floated.

From a start-up perspective, the amplifier, when the input voltage rampsup, could start up in a state where, if the output-common mode is low,for example near ground, then the input to the amplifier will also bepulled down to ground. That is, the input to the amplifier will besubstantially the same voltage as the output to the amplifier.Therefore, on start-up, it cannot be certain as to which voltage theamplifier will assume when it is initially powered up, since this cannotbe easily controlled. Therefore, if the output-common mode happens to bevery low, the input-common mode will also be very low, and that willshut off the amplifier.

In the illustrative embodiment of FIG. 6, the exemplary start-up circuit600 senses whether the aforementioned or a similar start-up problem hasmanifested itself, and then forces the output node of the amplifier 404to rail. That is, the start-up circuit 600 essentially rails it toV_(DD), therefore bringing the amplifier out of the bad start-upcondition, then turns the start-up circuit 600 off. After operation ofthe start-up circuit 600, the amplifier 404 will then assume a moresuitable state of operation. The nature of the amplifier 404 is suchthat if the output-common mode is too high, that is operating nearV_(DD) rail, then the amplifier 404 is still functional, and is devoidof voltage-related start-up problems. On the other hand, if theoutput-common mode of the amplifier 404 is too low, the amplifier 404will not start up.

In the illustration of FIG. 6, the differential input ports Vg+ and Vg−,the differential output ports Vo+ and Vo−, and voltage sources Vb1-Vb5of the amplifier 404 are shown. Source terminals of input differentialactive devices 601 and 602 form a common-source node (cmsrc). Alsoincluded in the embodiment of FIG. 6 is the exemplary start-up circuit600 and active devices 603-604.

The start up circuit 600 includes a comparing device, such as acomparator 606 and active devices 608 and 610. In the embodiment of FIG.6, the active devices 601-604, 608, and 610 are field effect transistors(FETs), although other active device types can be used. Common-modelatch-up, for example, can pull the differential output terminal Vo+ andVo− to ground. To counteract this effect, the comparator 606 monitors avoltage Vcmsrc at the cmsrc node to determine if this voltage goes belowa predetermined amount. If Vcmsrc falls below the predetermined amount,the comparator 606 produces an output compensatory voltage Vcmp to apositive supply level to pull up the level of Vo+ and Vo−. Thisultimately pulls Vcmsrc back up, as explained in greater detail below.

In monitoring Vcmsrc, the comparator 606 determines whether Vref isgreater than Vcmsrc. If Vref is greater, the comparator 606 outputs thecompensatory voltage Vcmp at positive supply. That is, if the positiveterminal of the comparator 606 is larger than its negative terminal, apositive supply voltage is produced as an output. This compensatoryoutput voltage Vcmp then turns on the devices 608 and 610.

When the devices 608 and 610 are activated by the compensatory voltageVcmp, they in-turn pull voltages Vd+ and Vd−, shown in FIG. 6, basicallyto ground. Vd+ and Vd− being pulled to ground temporarily turn off thedevices 603 and 604. This allows Vo+ and Vo− to be pulled back up. OnceVo+ and Vo− are pulled back up, Vg+ and Vg− are pulled up as wellthrough the resistive feedback network 400, shown in FIG. 4. Once Vg+and Vg− are pulled up, the active devices 601 and 602 begin conductingcurrent, enabling the amplifier 404 to reach a stable start-up state.Furthermore, once Vg+ and Vg− are pulled up, Vcmsrc is also pulled up.When Vcmsrc exceeds Vref, the compensatory output voltage Vcmp getspulled to ground so that the devices 608 and 610 are turned off.Consequently, the amplifier 404 returns to a normal operation mode.

FIG. 7 is an illustration of an exemplary method 700 of practicing thepresent invention. In FIG. 7, a comparing device in a systemfunctionally analogous to the system of FIG. 6 will compare thecommon-mode source voltage Vcmsrc to the reference voltage Vref in block702. As a result of the comparison, a compensating voltage Vcmp isproduced as an output to the comparing device, as depicted in block 704.Finally, the voltages Vo+ and Vo− are adjusted in accordance with thecompensating voltage Vcmp, as depicted in a block 706, and the amplifierthen returns to the normal operating state.

FIGS. 8A-8C are an illustration of a resistor layout constructed andarranged in accordance with another aspect of the present invention.More specifically, FIGS. 8A-8C depict a technique for geometricallypositioning resistors R1-R4 around the PGA to insure matching impedancesacross the associated IC. The layout technique can be used, for example,in construction of the resistive network 400 discussed above and can beimplemented using standard IC chip fabrication processes, materiel, andequipment.

The problem addressed by the resistor layout is that due to ICmanufacturing process variations, a mis-match between components, forexample resistors or resistances across the IC, may result. In ICmanufacturing, it is desirable that resistors of equal impedance matchsubstantially well across the entire substrate of the IC.

In the FIG. 8A, each of the resistors R1-R4 is representative of asingle resistor value. First, the value of each of the resistors R1-R4is split to form a number of corresponding respective resistor valuesR1′-R4′ as shown in FIG. 8B. By then configuring the resistors as shownin FIG. 8C (i.e., forming an interdigital structure across thesubstrate), substantially equal impedance values can be achievedthroughout all the resistors. That is, the geometric pattern establishedby the resistors R1′-R4′ connected along points A, B and C, and alongpoints D, E and F, of FIG. 8C provide a means of achieving substantiallyequal impedance values throughout all of the resistors.

The arrangement shown in FIGS. 8A-8C is a type of a common centroidlayout, including a technique of splitting resistance values amongmultiple resistors having intertwining paths. Although two seriesresistors are illustrated in FIG. 8B, in practice the number ofresistors can be extended to any suitable number needed to meetperformance requirements. Additionally, the arrangement illustrated inthe embodiment of FIG. 8B can also be extended to more parallel pathssnaking through the substrate and then recombining in an appropriatefashion as indicated in FIG. 8C.

In FIG. 8C, as noted, the resistors R1′ between points A and B form twoparallel paths. That is, a single path begins at point A, diverges, thenrecombines at point B. A feature of the present embodiment is that Nnumber of parallel paths can be split to cover different areas of asubstrate so that when they recombine, the device variation over thegeometry of the substrate cancels itself out. Therefore, the net resultof the technique of FIGS. 8A-8B is an interdigital device with averagedimpedances as opposed to a skewed device.

The foregoing description of the preferred embodiments provide anillustration and description, but is not intended to be exhaustive or tolimit the invention to the precise form disclosed. Modifications andvariations are possible consistent with the above teachings or may beacquired from practice of the invention. Thus, it is noted that thescope of the invention is defined by the claims and their equivalents.

What is claimed is:
 1. A method of reducing impedance variations in anelectrical circuit structured and arranged for placement on anintegrated circuit (IC) substrate, the electrical circuit including anelectrical component having (i) input and output ports and (ii) aplurality of cascaded impedance devices configured for connection alonga feedback path formed between the input and output ports, eachimpedance device having a predetermined impedance value, the methodcomprising: forming sets of parallel connected resistors, each set (i)corresponding to one of the impedance devices, (ii) including two ormore parallel resistor paths, each resistor path including two or morecascaded resistors, and (iii) having a total impedance valuesubstantially equal to the predetermined impedance value of itscorresponding impedance device; and configuring the sets of parallelresistor paths to form an interdigital structure across the substratewhen the electrical circuit is placed thereon, the interdigitalstructure being formed when the parallel resistor paths split at onepoint on the substrate and recombine at another point.
 2. An apparatusfor reducing impedance variations in an electrical circuit structuredand arranged for placement on an integrated circuit (IC) substrate, theelectrical circuit including an electrical component having (i) inputand output ports and (ii) a plurality of cascaded impedance devicesconfigured for connection along a feedback path formed between the inputand output ports, each impedance device having a predetermined impedancevalue, the apparatus comprising: means for forming sets of parallelconnected resistors, each set (i) corresponding to one of the impedancedevices, (ii) including two or more parallel resistor paths, eachresistor path including two or more cascaded resistors, and (iii) havinga total impedance value substantially equal to the predeterminedimpedance value of its corresponding impedance device; and means forconfiguring the sets of parallel resistor paths to form an interdigitalstructure across the substrate when the electrical circuit is placedthereon, the interdigital structure being formed when the parallelresistor paths split at one point on the substrate and recombine atanother point.
 3. The apparatus of claim 2, wherein the electricalcomponent is a programmable gain amplifier (PGA).
 4. The apparatus ofclaim 3, wherein the PGA is a differential amplifier.
 5. The apparatusof claim 4, wherein the predetermined impedance values of all of theimpedance devices are substantially equal.
 6. The apparatus of claim 5,wherein the IC is formed in CMOS.